OFDM is a transmission method according to which a large number of mutually orthogonal carriers are modulated with digital data to be transmitted and the resulting modulated waves are multiplexed and transmitted. With an increase in the number of carriers used, the symbol duration increases as compared with a single-carrier transmission method at the same transmission rate. This makes the OFDM transmission more robust to the effect of multi-path propagation. In OFDM, in addition, a redundant duration called a guard interval is inserted between adjacent symbols to avoid the inter-symbol interference that multi-path propagation would cause.
However, in a multi-path, the phases and amplitudes of the individual carries vary, so that the receiver needs to compensate (equalize) the distortion in phase and amplitude. In one scheme employed widely for the equalization, pilot signals (of which phase and amplitude are known to the receiver) are transmitted in some of the cells (i.e., the transmission units identified by combinations of the symbol number and the carrier number) contained in an OFDM signal. The receiver estimates the channel characteristics by using the pilot signals and equalizes the received signal by using the estimated channel characteristics.
For example, according to DVB-T (Digital Video Broadcasting Terrestrial) system, which is the standard for the broadcast transmission of digital terrestrial television in Europe, and ISDB-T (Integrated Services Digital Broadcasting Terrestrial) system in Japan, pilot signals called SPs (Scattered Pilots) are scattered on a carrier-symbol plane (hereinafter referred to as “k-n plane”) in a pattern shown in FIG. 17 (See Non-Patent Documents 1 and 2 listed below). In each figure showing a signal arrangement pattern of SP signals, the vertical axis represents a time axis and “n” represents a symbol number, whereas the horizontal axis represents a frequency axis, and k represents a carrier number. In addition, an open circle represents a cell carrying an SP signal, whereas a black dot represents a cell carrying data. Data referred herein includes data representing video and audio information and also include control information, such as TPS (Transmission Parameter Signaling) in DVB-T and TMCC (Transmission Multiplexing Configuration Control) in ISDB-T. In each figure showing a signal arrangement pattern of SP signals on the k-n plane, the symbol numbers starts from 0 and the carrier numbers starts from 0.
Here, let Tu represents the useful symbol duration, Tg represents the guard interval duration, and Ts (=Tu+Tg) represents the symbol duration. Then, the interval between adjacent cells in the same carrier in the direction of the time axis is equal to Ts, and the interval between adjacent cells in the same symbol in the direction of the frequency axis is equal to 1/Tu.
In FIG. 17, SP signals in each symbol appear at an interval of 12 carriers and SP signals in each carrier appear at an interval of 4 symbols. The position of each SP signal is shifted by three carriers per symbol. That is, when kSP(n) denotes the carrier number of a cell containing an SP signal in a symbol having the symbol number n, this carrier number kSP(n) satisfies the following Equation 1, where mod is a modulus operator and p is an integer greater than or equal to 0.kSP(n)=3×(n mod 4)+12×p  [Equation 1]
Each SP signal is modulated based on the pseudo-random binary sequence wk, and the amplitude and phase of the SP signal is determined depending exclusively on the carrier number k of the cell containing that SP signal and not on the symbol number n.
With reference to FIG. 18, the following describes the principles of the channel estimation and equalization performed by the receiver by using SP signals. FIG. 18 is a block diagram showing the structure of a typical receiver.
In a receiver 100, a signal received with a non-illustrated receiving antenna is subjected to predetermined processes by non-illustrated components including a tuner. The processed signal is supplied to a Fourier transform unit 101 where signal parts each containing a useful symbol duration Tu are clipped from the supplied signal and the Fourier transform is applied to the clipped signal parts to convert the clipped signal parts into a reception signal Y′(n, k). The converted reception signal Y′(n, k) is output to a division unit 106 and also to an SP extraction unit 102. The SP extraction unit 102 extracts a reception SP signal Y′ (n, kSP(n)) from the reception signal Y′ (n, k) and outputs the extracted reception SP signal Y′ (n, kSP(n)) to a division unit 104.
An SP generation unit 103 generates a nominal SP signal Y(n, kSP(n)), which is identical to an SP signal generated by the transmitter, and outputs the SP signal Y (n, kSP(n)) to the division unit 104. The division unit 104 divides the reception SP signal Y′ (n, kSP(n)) by the SP signal Y (n, kSP(n)) and outputs the division result as the channel response H′ (n, kSP(n)) to an interpolation unit 105. The interpolation unit 105 interpolates the channel response H′ (n, kSP(n)) on the k-n plane to estimate the channel response H′(n, k) for each cell and outputs the thus estimated channel response H′(n, k) to the division unit 106.
The division unit 106 divides the reception signal Y′(n, k) by the channel response H′(n, k) to estimate a transmission signal X′(n, k) and outputs the thus estimated transmission signal X′(n, k).
Through the above processes, the distortion in amplitude and phase of the transmission signal caused by multi-path is compensated using SP signals (See Patent Document 1, for example).
In addition, disclosed is the application of MIMO (Multiple Input Multiple Output) techniques, which employ multiple antennas at both the transmitter and receiver to achieve high-speed and high-capacity data transmission, to a digital terrestrial television broadcasting using SP signals, such as DVB-T (See Non-Patent Document 3, for example).
First, the following describes the overview of a MIMO transmission system in which the transmitter and the receiver both have two antennas, with reference to FIG. 19. FIG. 19 is a diagram showing such a MIMO transmission system.
A transmitter 200 transmits a first transmission signal and a second transmission signal from a first transmitting antenna 201 and a second transmitting antenna 202, respectively. The first transmission signal is obtained by applying the inverse Fourier transform to a first transmission signal Xc1(n, k), and the second transmission signal is obtained by applying the inverse Fourier transform to a second transmission signal Xc2 (n, k). Note that the first and second transmission signals are simultaneously transmitted respectively on the cells each having the symbol number n and the carrier number k.
A receiver 300 receives a first reception signal with a receiving antenna 301. The first reception signal contains the first transmission signal arrived via a channel Pc11 and the second transmission arrived via a channel Pc12. The receiver 300 applies the Fourier transform to the first reception signal to obtain a first reception signal Yc′1(n, k). In addition, the receiver 300 receives a second reception signal with a receiving antenna 302. The second reception signal contains the second transmission signal arrived via a channel Pc22. The receiver 300 applies the Fourier transform to the second reception signal to obtain a second reception signal Yc′2 (n, k). The receiver 300 then conducts a predetermined process on the first reception signal Yc′1(n, k) and the second reception signal Yc′2(n, k) and outputs the first transmission signal Xc′1(n, k) and the second transmission signal Xc′2(n, k).
Here, let Hc11(n, k), Hc12(n, k), Hc21(n, k), and Hc22 (n, k) respectively denote the channel responses of channels Pc11, Pc12, Pc21, and Pc22 at the cell having the symbol number n and the carrier number k. Let Nc1(n, k) and Nc2(n, k) denote the noise power contained in the first reception signal Yc′1(n, k) and in the second reception signal Yc′2(n, k), respectively. Then, the first reception signal Yc′1(n, k) and the second reception signal Yc′2(n, k) are expressed by Equation 2 shown below. The notation [ ] in Equation 2 represents a matrix.
                              [                                                                                          Yc                    ′                                    ⁢                  1                  ⁢                                      (                                          n                      ,                      k                                        )                                                                                                                                            Yc                    ′                                    ⁢                  2                  ⁢                                      (                                          n                      ,                      k                                        )                                                                                ]                =                              [                                                                                Hc                    ⁢                                                                                  ⁢                    11                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                                            Hc                    ⁢                                                                                  ⁢                    12                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                                                                        Hc                    ⁢                                                                                  ⁢                    21                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                                            Hc                    ⁢                                                                                  ⁢                    22                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                            ]                    ⁢                                                                 [                                                                                                    Xc                        ⁢                                                                                                  ⁢                        1                        ⁢                                                  (                                                      n                            ,                            k                                                    )                                                                                                                                                                        Xc                        ⁢                                                                                                  ⁢                        2                        ⁢                                                  (                                                      n                            ,                            k                                                    )                                                                                                                    ]                            +                              [                                                                                                    Nc                        ⁢                                                                                                  ⁢                        1                        ⁢                                                  (                                                      n                            ,                            k                                                    )                                                                                                                                                                        Nc                        ⁢                                                                                                  ⁢                        2                        ⁢                                                  (                                                      n                            ,                            k                                                    )                                                                                                                    ]                                                                        [                  Equation          ⁢                                          ⁢          2                ]            
That is, once the channel responses of the channels Pc11, Pc12, Pc21, and Pc22 are estimated, the receiver 300 is able to separate and equalize the first transmission signal Xc′1 (n, k) and the second transmission signal Xc′ 2(n, k) by using Equation 3 shown below, where Hc′11(n, k), Hc′12(n, k), Hc′21(n, k), and Hc′22(n, k) are the channel responses estimated by the receiver 300. In Equation 3, the notation [ ] represents a matrix, and the notation [ ]−1 represents the inverse matrix of [ ].
                              [                                                                                          Xc                    ′                                    ⁢                  1                  ⁢                                      (                                          n                      ,                      k                                        )                                                                                                                                            Xc                    ′                                    ⁢                  2                  ⁢                                      (                                          n                      ,                      k                                        )                                                                                ]                =                                            [                                                                                                                  Hc                        ′                                            ⁢                                                                                          ⁢                      11                      ⁢                                              (                                                  n                          ,                          k                                                )                                                                                                                                                Hc                        ′                                            ⁢                                                                                          ⁢                      12                      ⁢                                              (                                                  n                          ,                          k                                                )                                                                                                                                                                                Hc                        ′                                            ⁢                                                                                          ⁢                      21                      ⁢                                              (                                                  n                          ,                          k                                                )                                                                                                                                                Hc                        ′                                            ⁢                                                                                          ⁢                      22                      ⁢                                              (                                                  n                          ,                          k                                                )                                                                                                        ]                                      -              1                                ⁡                      [                                                                                                      Yc                      ′                                        ⁢                    1                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                                                                                              Yc                      ′                                        ⁢                    2                    ⁢                                          (                                              n                        ,                        k                                            )                                                                                            ]                                              [                  Equation          ⁢                                          ⁢          3                ]            
Non-Patent Document 3 describes a technique for enabling separation and estimation of channel responses of two channels from two transmitting antennas to one receiving antenna, by transmitting SP signals arranged in the pattern shown in FIG. 17 from the first transmitting antenna and SP signals arranged in the pattern shown in FIG. 20 from the second transmitting antenna. In FIG. 20, a plus (+) sign indicates that the polarity of an SP signal transmitted from the second transmitting antenna is not inverted with respect to the polarity of a corresponding SP signal transmitted from the first transmitting antenna. On the other hand, a minus (−) sign indicates that the polarity of an SP signal transmitted from the second transmitting antenna is inverted with respect to the polarity of a corresponding SP signal transmitted from the first transmitting antenna.
That is, of the SP signals transmitted from the second transmitting antenna, the polarity of each SP signal having an even symbol number is not inverted and of each SP signal having an odd symbol number is inverted, with respect to the polarity of a corresponding SP signal transmitted from the first transmitting antenna.
The receiver observes, for each symbol where the symbol number n is an even number, components representing the sum of the channel responses of the two channels, one of which is from the first transmitting antenna to the receiving antenna and the other is from the second transmitting antenna to the receiving antenna (hereinafter, the former is referred to as “first channel response” and the latter as “second channel response”). On the other hand, for each symbol where the symbol number n is an odd number, components representing the difference between the first and second channel responses are observed. Therefore, the receiver can separate and estimate the first channel response by adding the sum components and the difference components, and the second channel response by subtracting the difference components from the sum components.
[Non-Patent Document 1]
    “Digital Video Broadcasting (DVB); Framing structure, Channel coding and modulation for digital terrestrial television”, ETSI EN 300 744 by European Telecommunications Standards Institutes[Non-Patent Document 2]    “TRANSMISSION SYSTEM FOR DIGITAL TERRESTRIAL TELEVISION BROADCASTING”, ARIB STD-B31 by Association of Radio Industries and Businesses[Non-Patent Document 3]    “A DUAL POLARIZATION MIMO BROADCAST TV SYSTEM”, BBC Research White Paper WHP 144 by J. D. Mitchell, P. N. Moss and M. J. Thorp[Patent Document 1]    JP patent No. 2772286
The following now considers the range in which a channel response is duly estimated on condition that SP signals are arranged in the pattern shown in FIG. 17, which is used in the DVB-T system as well as in the ISDB-T system.
FIG. 21 is a schematic view of responses on the delay time-Doppler frequency plane (hereinafter referred to as the “τ-fD plane”) of SP signals arranged on the k-n plane in the pattern shown in FIG. 17. In other words, FIG. 21 show two-dimensional Fourier transform pairs of SP signals arranged on the k-n plane in the pattern shown in FIG. 17. In each figure showing SP signal responses and showing the estimatable ranges of channel responses, the horizontal axis represents a delay time axis (hereinafter referred to as the “τ axis”) and corresponds to the delay time (T) of the impulse response of a channel. The vertical axis represents a Doppler frequency axis (hereinafter referred to as the “fD axis”) and corresponds to the Doppler frequency (fD) of the Doppler spectrum of a channel. In addition, a black dot represents a response of an SP signal on the τ-fD plane.
As shown in FIG. 21, the minimum interval between SP signal responses on the τ-fD plane in the τ axis direction is equal to Tu/12. It is because SP signals on the k-n plane are arranged to appear one for every 12 carriers in the same symbol. In other words, the sampling interval in the k axis direction is equal to 12/Tu. Further, the minimum interval between SP signal responses on the τ-fD plane in the fD axis direction is equal to 1/(4Ts). It is because SP signals on the k-n plane are arranged to appear one for every 4 symbols in the same carrier. In other words, the sampling interval in the n axis direction is equal to 4Ts. Still further, the minimum interval between SP signal responses on the τ-fD plane at the same Doppler frequency in the τ axis direction is equal to Tu/3. It is because the minimum interval between SP signals on the k-n plane in the k axis direction is equal to 3 carriers. Still further, the minimum interval between SP signal responses at the same delay time on the τ-fD plane in the fD axis direction is equal to 1/Ts. It is because the minimum interval between SP signals on the k-n plane in the n axis direction is equal to one symbol.
In the case where an impulse response of a channel has a delay spread, the response spreads in the τ axis direction as compared with a corresponding SP signal response. In the case where a Doppler spectrum of a channel has a frequency spread, the spectrum spreads in the fD axis direction as compared with a corresponding SP signal response.
FIG. 22 shows a region of the τ-fD plane in which the channel response H′ (n, kSP(n)) of an SP signal can be interpolated without causing aliasing distortion, on condition that the channel response H′ (n, kSP(n)) is first interpolated in the n axis direction and then in the k axis direction of the k-n plane. In FIG. 22, a black dot represents an SP signal response on the τ-fD plane, and a rectangle represents a channel response of the channel from the transmitting antenna to the receiving antenna.
From FIG. 22, it is known that a rectangular region having a width of Tu/3 in the τ axis direction and a width of 1/(4Ts) in the fD axis direction is the region in which the channel response is interoperated without causing aliasing distortion (hereinafter, referred to as “interpolatable region”). According to the DVB-T and ISDB-T systems, the length of the longest guard interval duration is Tu/4. With the guard interval duration equal to Tu/4, the spread of the impulse response of the channel equal to Tu/4 or less would not adversely affect the reception quality. It is because the inter-symbol interference is ensured to fall within the guard interval duration. The width of the interpolatable region in the τ axis direction is set to Tu/3 in order to allow a margin for practical filters and yet to ensure a correct estimation of a channel response without incurring the risk of inter-symbol interference.
As described above, in terms of the design details of a transmission system, the guard interval duration and the SP signal arrangement are closely related. That is, in order not to impair the tolerance to multi-path delay provided by insertion of guard interval durations, the minimum interval between SP signals on the k-n plane in the k axis direction needs to be shorter than a predetermined interval. In terms of the transmission efficiency, however, it is desirable to keep to a minimum the density of SP signals, which do not carry any useful information. That is, there is a trade-off between the guard interval duration and the SP signal arrangement.
FIG. 23 shows a region of the τ-fD plane in which the channel response H′ (n, kSP(n)) of an SP signal can be interpolated without causing aliasing distortion, on condition that the channel response H′ (n, kSP(n)) is interpolated only in the k axis direction and not in the n axis direction of the k-n plane. In FIG. 23, a black dot represents an SP signal response on the τ-fD plane, and a rectangle represents a channel response of the channel from the transmitting antenna to the receiving antenna.
From FIG. 23, it is known that a rectangular region having a width of Tu/12 in the τ axis direction and a width of 1/Ts in the fD axis direction is a region in which the channel response is interoperated without causing aliasing distortion (hereinafter, referred to as “interpolatable region”).
The following now considers the range in which channel response is duly estimated with the SP signal arrangement disclosed in Non-Patent Document 3, which is used for a MIMO transmission system.
The process of inverting and not inverting the polarity of SP signals transmitted from the first transmitting antenna is equivalent to an arithmetic operation of multiplying individual SP signals transmitted from the first transmitting antenna, by the complex plane wave expressed by the left side of Equation 4 shown below. The complex plane wave has an equi-phase line parallel to the k axis direction on the k-n plane, and the cycle in the n axis direction is equal to 2n.
                              exp          ⁡                      (                          j2π              ⁢                              1                2                            ⁢              n                        )                          =                  exp          ⁡                      (                          j2π              ⁢                              1                                  2                  ⁢                  Ts                                            ⁢              t                        )                                              [                  Equation          ⁢                                          ⁢          4                ]            
Note that in Equation 4, the right side is obtained by rewriting the left side using the relation n=(1/Ts)t.
Accordingly, the response of each SP signal transmitted from the second transmitting antenna is said to be shifted the response of a corresponding SP signal transmitted from the first transmitting antenna, by 1/(2Ts) in the fD axis direction on the τ-fD plane.
In view of the above, the responses of SP signals transmitted from the first transmitting antenna and the responses of SP signals from the second transmitting antenna are expressed on the same τ-fD plane as shown in FIG. 24. Note that a black dot represents a response of an SP signal transmitted from the first transmitting antenna, whereas a cross represents a response of an SP signal transmitted from the second transmitting antenna.
Note that the process of inverting and not inverting the polarity of SP signals transmitted from the first transmitting antenna shown in FIG. 20 is to invert the polarity of an SP signal transmitted from the first transmitting antenna at every third carrier in the frequency direction. In other words, the process may be construed to be equivalent to an arithmetic operation of multiplying individual SP signals transmitted from the first transmitting antenna, by the complex plane wave expressed by the left side of Equation 5 shown below. The complex plane wave has an equi-phase line parallel to the n axis on the k-n plane and the cycle in the k axis direction is equal to 6k.
                              exp          ⁡                      (                                          -                j2π                            ⁢                              1                6                            ⁢              k                        )                          =                  exp          ⁡                      (                                          -                j2π                            ⁢                              Tu                6                            ⁢              f                        )                                              [                  Equation          ⁢                                          ⁢          5                ]            
Note that in Equation 5, the right side is obtained by rewriting the left side using the relation k=Tuf. In addition, the phase term in Equation 5 is attached with a negative (−) sign. It is because the delay in the positive direction along the τ axis corresponds to the phase rotation exp(−j2πfτ) in the negative direction in proportion to the frequency f.
Based on the above understanding, it is said that the response of each SP signal transmitted from the second transmitting antenna is shifted the response of a corresponding SP signal transmitted from the first transmitting antenna, by Tu/6 in the τ axis direction on the τ-fD plane. It is thus apparent from that each response shown in FIG. 20 is equivalent to that obtained by shifting the response of a corresponding SP signal shown in FIG. 24 by 1/(2Ts) in the fD axis direction.
The receiver divides each received SP signal (i.e., a mixed SP signal which is a mixture of an SP signal transmitted from the first transmitting antenna and an SP signal transmitted from the second transmitting antenna) by the nominal SP signal. As a result of the division, the receiver obtains a channel response which is a mixture of a channel response of the channel from the first transmitting antenna to the receiving antenna (the first channel response) and a channel response of the channel from the second transmitting antenna to the receiving antenna (the second channel response).
The first channel response has the spreading from the black dots shown in FIG. 24, in accordance with the impulse response and Doppler spectrum. Similarly, the second channel response has the spreading from the crosses shown in FIG. 24, in accordance with the impulse response and Doppler spectrum.
FIG. 25 shows a region of the τ-fD plane in which the first and second channel responses are interpolated without causing aliasing distortion and separated from each other without causing crosstalk therebetween, on condition that the channel response of each SP signal is interpolated first in the n axis direction and then in the k axis direction on the k-n plane. In FIG. 25, a black dot represents a response of an SP signal transmitted from the first transmitting antenna, whereas a cross represents a response of an SP signal transmitted from the second transmitting antenna. In addition, a rectangular with a solid line represents the first channel response, whereas a rectangular with a broken line represents the second channel response.
From FIG. 25, it is known that a rectangular region having a width of Tu/6 in the τ axis direction and a width of 1/(4Ts) in the fD axis direction is what is hereinafter referred to as “interpolatable & separable region”. In the interpolatable & separable region, the first and second channel responses are interoperated without causing aliasing distortion and separated without causing crosstalk therebetween.
FIG. 26 shows a region of the τ-fD plane in which the first and second channel responses are interpolated without causing aliasing distortion and separated from each other without causing crosstalk therebetween, on condition that the channel response of each SP signal is interpolated in the k axis direction only and not in the n axis direction on the k-n plane. In FIG. 26, a black dot represents a response of an SP signal transmitted from the first transmitting antenna, whereas a cross represents a response of an SP signal transmitted from the second transmitting antenna. In addition, a rectangular with a solid line represents the first channel response, whereas a rectangular with a broken line represents the second channel response.
From FIG. 26, it is known that a rectangular region having a width of Tu/12 in the τ axis direction and a width of 1/(2Ts) in the fD axis direction is what is hereinafter, referred to as “interpolatable & separable region”. In the interpolatable & separable region, the first and second channel responses are interoperated without causing aliasing distortion and separated without causing crosstalk therebetween.
From a comparison of the interpolatable region shown in FIG. 22 with the interpolatable & separable region shown in FIG. 25, it is shown that the width Tu/6 of the interpolatable & separable region in the τ axis direction is a half of the width Tu/3 of the interpolatable region in the τ axis direction. As mentioned above, it is preferable to set the τ-axis direction width in which correct estimation of the first and second channel responses is ensured in a manner not to impair the tolerance to multi-path delay provided by insertion of guard interval durations. However, the SP signal transmission method described in Non-Patent Document 3 is associated with the following problem, even without considering any margin to be allowed for practical filters used for interpolation and separation. That is, in the case where the guard interval duration is longer than Tu/6, specifically where the guard interval duration is equal to Tu/4 for example, the tolerance to multi-path delay achieved by the insertion of guard intervals is impaired and thus the first and second channel responses may not be correctly estimated.
In addition, from a comparison of the interpolatable region shown in FIG. 23 with the interpolatable & separable region shown in FIG. 26, it is shown that the width 1/(2Ts) of the interpolatable & separable region in the fD axis direction is a half of the width 1/Ts of the interpolatable region in the fD axis direction. As clarified above, the SP signal transmission method according to Non-Patent Document 3 has a problem in the ability of following the time variation of a channel.
In view of the problems noted above, the present invention aims to provide a transmitter, a receiver, and an OFDM transmission method each of which achieves the following advantages, in the case where a plurality of pilot signals are transmitted from a plurality of transmitting antennas. The transmitter, receiver, and OFDM transmission method according to the present invention ensure correct estimation of a channel response involving a delay spread to the comparable to the case where pilot signals are transmitted from a single transmitting antenna or ensure the ability to follow the time variation of a channel to the extent comparable to the case where pilot signals are transmitted from a single transmitting antenna.